Compact, Wideband CPW-to-Slotline Multimodal Transition

Marco Antonio Llamas1, David Girbau1, Miquel Ribó2, Lluís Pradell1 1Signal Theory and Communications Department, Universitat Politècnica de Catalunya

1-3 Jordi Girona, 08034 Barcelona, Spain. Phone: +34-93-401.68.34

pradell@tsc.upc.edu 2Signal Theory and Communications Department - Enginyeria La Salle - Universitat Ramon Llull (URL)

Pg. Bonanova 8 - 08022 Barcelona, Spain

Abstract—. This paper presents and implements a number of configurations for compact, wideband, low-loss, uniplanar coplanar (CPW)-to-Slotline transitions. These are modelled using a multimodal model and it is theoretically and empirically proved what are the main design parameters and their influence on the final transition performance. An optimized transition topology is implemented on alumina, featuring a state-of-the-art maximum insertion loss lower than 0.7 dB in the frequency band from DC to 50 GHz, with a reduced area of 0.105 mm2.

I. INTRODUCTION CPW-to-Slotline or CPW-to-Coplanar strip (CPS)

transitions are necessary to implement a number of applications in uniplanar technology such as baluns for frequency doublers [1] or push-pull amplifiers [2], mixers [3]-[5], and interconnections between balanced structures and tapered slot antennas [6]-[7]. These transitions are also used in alternative structures with wider bandwidth than conventional structures, as shown in [8].

Many transition topologies have been proposed in the literature in order to obtain low-loss and wideband performance. Reference [9] takes advantage of the Marchand balun configuration synthesized in uniplanar technology to build broadband transitions, achieving a 1 dB insertion loss in the frequency band from 1.48 to 6.02 GHz in a back-to-back configuration. References [10] and [11] present a double Y transition topology, achieving 1 dB insertion loss in the frequency band from 3 to 21 GHz in a back-to-back configuration [11]. The disadvantage of the double Y topology is that the CPW signal line and one of its ground planes are interconnected. This sometimes can be a problem, as for instance in some mixers, where there is a need for DC isolation, since it does not permit biasing of shunt mounted devices. In [12], different versions of the double Y topology and Marchand balun-based transitions are presented. Some of the compensation stubs are integrated inside the central CPW line in order to reduce transition dimensions. Another transition topology is presented in [13], where the CPW even mode is converted into a Slotline mode by reversing the phase of the wave propagating along one of the CPW slots with respect to the other. This transition features (in a back-to-back configuration) a 3dB insertion loss in the frequency band from 10.5 to 18 GHz. In [14] the same transition type is used,

integrating a slow-wave structure in order to reduce the dimensions of the transition presented in [13] up to 50%. Electrical lengths associated with open-ended and short- ended stubs restrict the compactness and bandwidth of the above transitions. Design equations for double Y and Marchand balun-based transitions are given in [15].

Finally, other transition types which are easier to compact and exhibit wider bandwidth than the referenced above are presented in the literature [16]-[17]. Tuning structures are not used in this type of transitions. The transition presented in [16] (see Fig. 1a), features a maximum insertion loss of 1 dB in the frequency range from DC to 26 GHz in a back-to-back configuration. A model for this last topology of transitions is presented in [18], which is based on the multimodal behaviour of the lines used in the transition; it permits to quantify the conversion between the different modes present at the transition plane (CPW even mode, CPW odd mode and slotline mode). To the best of our knowledge, there is not any systematic study in the literature of this kind of wideband transitions, as a function of its design parameters.

This paper presents and implements a number of configurations for wideband low-loss uniplanar CPW-to-Slotline transitions. These are modelled using the multimodal model proposed in [18] and their parameters optimized according to this theoretical analysis. The implementation of the proposed transitions on an alumina substrate shows a low loss (1.4 dB), broadband behaviour up to 50 GHz in a back-to-back configuration.

II. STRUCTURE AND MODEL

The selected configuration for this study is a modification of the transitions presented in [16]-[17]. It can be considered as a transition between a finite ground plane CPW (CPWFGP), propagating the even mode, and a CPS propagating the slotline mode. In this configuration, one slot of the CPWFGP is terminated with an open circuit and the other one is directly connected to the CPS slot. In order to terminate one of the CPWFGP slots in open circuit, references [16]-[17] connect the CPWFGP and the CPS in a right angle (see Fig. 1a). Thus the open circuit can be considered as a radial stub (loaded with an open circuit), which is susceptible to produce

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radiation losses. Another way to terminate one of the slots of the CPWFGP with an open circuit is to use an asymmetrical CPS, where one of its strips has the same width as the central CPWFGP strip [17] (see Fig. 1b), allowing an in-line configuration. This configuration can be more difficult to match and losses can increase due to the narrow strip that implements the CPS.

Fig.1. a/ transition between CPWFGP and CPS [16], b/ transition between CPWFGP and asymmetrical CPS (ACPS) [17].

Fig.2. Proposed Transition between finite ground planes CPW (CPWFGP) and symmetrical coplanar strip line (CPS).

The transition proposed in this work is shown in Fig.2. The

advantages of this topology are: direct connection between one of the CPW slots and the CPS slot; the characteristic impedance modification of the CPS can be controlled by enlarging or narrowing its strips; symmetrical CPS can be used to control the transition matching; tuned structures are not used (therefore the operation bandwidth is increased) and the in-line configuration permits maximum miniaturization; since the CPW signal line and its ground planes are isolated, shunt mounted devices can be biased easily.

In the general circuit multimodal model for CPW-to-Slotline transitions presented in [18], one of the CPW line slot (slot A) is loaded with arbitrary impedance ZA. The other slot (slot B) is directly connected to the slotline. According to this model, a matrix with the conversion ratios between the different modes at the transition plane as a function of ZA is extracted. For the transition presented in this work, the general model presented in [18] is particularized with ZA ∞, since the slot A is terminated with an open circuit. Then the circuit model proposed in [18] can be simplified to the one shown in Fig.3.

Fig. 3 Equivalent circuit model for the CPW-to-Slotline transition loaded with an open circuit.

Zoe and Zoo are the characteristic impedances of the CPW propagating the CPW even mode and the odd mode, respectively, and Zs is the characteristic impedance of the slotline. According to this model the [S] matrix, referenced to the characteristic impedances previously defined, which describes the modal transformation at the transition plane can be extracted:

1 8 41 8 1 4 .9

4 4 7

s s

e e

o o

b ab ab a

= −

−

(1)

In (1), as and bs are the incident and reflected waves of the

slotline mode, respectively, ae and be correspond to the CPW even mode, and ao and bo to the CPW odd mode. It can be deduced from (1) that the most important conversion ratio between modes take place between the slotline and CPW even mode (S21=8/9) when Zoe=Zoo=Zs.

This modal conversion between the CPW even mode and the slotline mode can be optimized by connecting an air bridge between the CPW lateral ground planes at the transition plane in order to avoid CPW odd mode propagation. Applying this condition the new circuital model is the one shown in Fig. 4.

Fig. 4 Equivalent circuit model for the CPW-to-Slotline transition loaded with an open circuit and with an air bridge between CPW ground planes.

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The new conversion ratios extracted from the circuital model presented in Fig. 4 are given in (2):

0 1 01 0 0 .0 0 0

s s

e e

o o

b ab ab a

=

(2)

It can be deduced from (2), that the modal transformation is

perfect between the CPW even mode and the slotline mode when Zoe=Zs.

It can be concluded from the previous analysis that the two most important parameters to achieve a low-loss broadband transition for these transitions are: impedance matching between CPW even mode and slotline mode, and the value of the load connected to the slot A of the CPW (ZA)..

III. IMPLEMENTATION AND EXPERIMENTAL RESULTS This section is devoted to empirically determine the

influence of the two design parameters analysed in last section (impedance matching between CPW even mode and slotline mode, and the ZA load) in the transition final performance. The study relies on modifying the impedance matching and the load ZA in the transition topology shown in Fig. 2, which has been proposed in order to minimize radiation losses produced by the open circuit termination and permits the use of symmetrical CPS. In order to load one of the CPW slots with an open circuit, a cut in one of the CPW lateral ground plane is made. Different transitions types (addressed as type I, II and III) have been implemented. The structures were patterned on a 254 µm thick alumina substrate (relative permittivity εr= 9.8, loss tangent tgδ=2·10-4) with a 5 µm-thick gold layer. The air bridges that connect the CPW lateral ground planes have been built with 17 µm-diameter, gold wire-bondings. CPWFGP line dimensions are (Fig. 2) w=50 µm and s=30 µm (resulting in an even mode characteristic impedance of 50 Ω), and the CPS slot width is s1=30 µm. The CPS ground plane width is obtained from electromagnetic simulation with ADS-MomentumTM in order to fit the required CPS characteristic impedance.

Fig. 5 shows the measured return and insertion loss from DC to 50 GHz for the type I transition in a back-to-back configuration, where the CPS characteristic impedance (Zcps) is 30 Ω. In order to synthesize the CPS impedance without changing the slot width, the strip widths were increased (1000 µm). In this case ZA behaves as an open circuit at low frequencies, but the quality of the open decreases as the frequency increases. The electrical length of the stub loaded with the open circuit becomes λ/4 around 30 GHz, being its input impedance a short circuit, quickly degrading the transition performance. Fig. 6 shows the measured return and insertion loss for the type II transition in a back-to-back configuration, with a Zcps of 44 Ω. In this case the CPS strip width is narrower than in type I (500 µm). Here the good quality of the open circuit is kept up to 40 GHz. Finally, Fig. 7 shows measured and simulated return and insertion loss for the type III transition in a back-to-back configuration, with a Zcps of 75 Ω. In this case the CPS strip width (100 µm) is still

narrower than in types I and II, keeping the open circuit good quality beyond the measurement range.

5 10 15 20 25 30 35 40 450 50

-30

-20

-10

-40

0

-20

-15

-10

-5

-25

0

freq, GHz

((

,)) (

())

Fig.5. Measured return and insertion loss of the type I transition in back-to-back configuration with Zcps=30 Ω.

5 10 15 20 25 30 35 40 450 50

-40

-30

-20

-10

-50

0

-8

-6

-4

-2

-10

0

freq, GHz

Fig.6 Measured return and insertion loss of the type II transition in back-to-back configuration with Zcps=44 Ω.

5 10 15 20 25 30 35 40 450 50

-40

-30

-20

-10

-50

0

-1.0

-0.5

-1.5

0.0

freq, GHz

dB(S

(1,1

)) dB(S

(2,1))

Fig.7. Simulated (symbol lines) and measured (continuous line) return an insertion loss of the type III transition in back-to-back configuration with Zcps=75 Ω.

Type III transition (see Fig. 7) achieves the best bandwidth

as already expected from numerical simulations. The measured results (Fig. 5-7) demonstrate that the quality of the open circuit plays a more important role than the impedance matching between CPWFGP and CPS lines in order to determine the transition performance.

Note that the structures presented in this study are in a back-to-back configuration in order to ease its on-wafer characterization by means of ground-signal-ground (GSG) probes. As these structures are symmetrical and reciprocal, the insertion loss of a single transition can be approximated to half of the back-to-back measured result. Therefore, transition type III shows an insertion loss lower than 0.7 dB in a

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wideband frequency range, from DC to 50 GHz, with a reduced area of 300 x 350 µm2.

CONCLUSIONS In this work, a general CPW-to-Slotline multimodal model is applied to the design of CPWFGP-CPS transitions in which one of the CPW slots is loaded with an open circuit. It is theoretically and empirically proved that the open circuit is a critical design parameter in the transition performance. The quality of the open circuit plays a more important role than the impedance matching between CPWFGP and CPS lines. Finally, an improved transition topology is implemented on alumina, featuring a state-of-the-art maximum insertion loss lower than 0.7 dB in the frequency band from DC to 50 GHz, with an area of 0.105 mm2.

ACKNOWLEDGEMENT This work has been supported by the Spanish Government

Project ESP2004-07067-C03-03.

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